1. Field of the Invention
The invention relates to the field of microwave transistor power amplifiers in a series configuration using a local feedback network to share power among a plurality of transistors for increased output power.
2. Description of the Prior Art
Traditionally, the microwave power amplifiers are designed in common source or common gate configuration, with the device size being determined by the required output power. Large device sizes ("device periphery") give large drive currents, but with invariant maximum device voltages with output power as the product of these two factors. With the voltage swing at the drain of the field effect transistor, FET, limited by the breakdown voltage of the transistor, obtaining large output powers results in very low and unfeasible values of load resistor.
The output power levels obtained from solid state microwave sources have been limited by a combination of factors involving semiconductor device physics, transmission line losses, and circuit implementation. These factors in combination lead to a maximum output power varying as the inverse square of the operating frequency, referred to as the Pf.sup.2 limit. The Pf.sup.2 limit arises through the combination of low breakdown voltages for high frequency transistors and high skin effect losses from low impedance transmission lines. In any microwave transistor, whether a MEtal Semiconductor Field Effect Transistor (MESFET), a High Electron Mobility Transistor (HEMT) or a bipolar transistor, there are two electron transport regions, a control region and a high field drift region. In MESFETs the control region is under the gate electrode, while the high field drift region is the lateral extension of the depletion layer between gate and drain.
To obtain a high current gain cutoff frequency essential for high frequency operation, the transit time for the drift region must be small. This transit time is given by .tau..sub.drift =D.sub.drift /2.nu..sub.sat, where D.sub.drift is the length of the drift region and .nu..sub.sat, the electron velocity. Since the current gain cutoff frequency, f.sub..tau., is inversely proportional to this transit time, EQU f.sub..tau. =.nu..sub.sat/.pi.D.sub.drift
assuming that drift region dominates the total transit time.
The drift region serves to support the transistor's output voltage, and can support some maximum field E.sub.max. The transistor breakdown voltage, V.sub.br, is the product of this maximum field, E.sub.max, and the thickness of the layer, V.sub.br =E.sub.max D.sub.drift. As a result, transistors designed for high breakdown voltage have thick drift regions with long electron transit times, hence low current gain cutoff frequencies. The breakdown voltage varies inversely with the current gain cutoff frequency, V.sub.br .varies.E.sub.max .nu..sub.sat (f.sub..tau.).sup.-1, and consequently high frequency transistors have low breakdown voltages.
If the transistor is operated into some load resistance R.sub.L, then the maximum power level which can be delivered to the load resistance is P.sub.max =V.sub.br.sup.2 /8 R.sub.L. Consequently, P.sub.max .varies.1/f.sup.2, where f is the operating frequency. This is the Pf.sup.2 limit.
The Pf.sup.2 limit exists only because a fixed minimum load resistance exists. As we scale the FET gate width, the transistor's available drive current increases proportionally. With a maximum peak-to-peak voltage swing of V.sub.br, the FET gate width is chosen so that the transistor current I.sub.dss, which is the drain-to-source saturation current, is the required peak-to-peak output current, V.sub.br /R.sub.L. The maximum available output power from the transistor is V.sub.br I.sub.dss /8, which increases without limit as we increase the FET gate width. But, with a minimum load resistance R.sub.L, further increases in I.sub.dss beyond V.sub.br /R.sub.L do not increase the output power beyond V.sub.br.sup.2 /8R.sub.L, and the excess available transistor drive current is wasted.
Low values of load resistance R.sub.L are a major difficulty because of several factors. First, very low load resistances are not compatible with microwave connectors, transmission lines, microwave antennas, or other components, all of which are designed to operate into a standard 50 ohm interface impedance. Secondly, and more fundamentally, low circuit resistances result in substantial microwave power losses through skin effect resistances in the microwave connections, losses which become progressively more severe as the circuit impedance levels are reduced.
There are several standard microwave circuit design techniques which only partly alleviate this difficulty. At low frequencies, wire wound transformers are used to obtain a low load resistance. In an equivalent approach, at higher frequencies output reactive impedance matching networks are used to obtain a low transistor load impedance. The problems with these approaches are as follows. First, wire wound transformers do not function well at microwave frequencies, and impedance matching networks must instead be used. Secondly, matching networks of high impedance transformation ratios are hard to realize, are very sensitive to device parameters, and do not work over a broad frequency range. Third, the impedance matching networks themselves are transmission line systems with skin effect losses, and low transformed load resistances are obtained only at the penalty of significant skin effect power losses within the impedance matching network. The conclusion is that impedance transforming networks increase the attainable saturated power to a significant factor, but that there are maximum feasible transformation ratios which place limits on the attainable improvement in the saturated output power.
To see this concretely, consider a typical design problem. A GaAs power HEMT with transistor bandwidth sufficient for S-band operation will have .about.10 Volts breakdown. If 10 watts saturated output power is required, the resulting required load resistance (calculated from P.sub.max =V.sub.br.sup.2 /8R.sub.L) is 1.25 ohm. An impedance matching network transforming 1.25 ohm to 50 ohm will have a fractional bandwidth on the order of 2 to 5 percent, will be very big, and will have very high skin effect losses.
A third standard microwave power design technique is the use of power combining networks. One designs a transistor power amplifier stage with the highest feasible saturated output power. Taking two such power amplifiers, the power from an input signal source is divided equally between the two transistors using a 2:1 power dividing network (e.g. a Wilkinson or Lange Coupler). The output powers of the two transistors are recombined in a second Wilkinson or Lange power combiner, and the power delivered to the load. In this manner the saturated output power is increased 2:1. The saturated output power can be increased 2:1, 4:1, 8:1, 16:1, etc., by repeating the power combining technique in multiple steps. The limitations to such "corporate" power combining networks are similar to those of impedance matching networks for microwave power amplifier designs, namely as the level of power combining is increased, the physical size of the power combiners increases, their skin effect losses increase, and the feasible system bandwidth decreases.
The above summarizes the state of the art in microwave power amplifier design. Typical microwave power amplifiers use a combination of impedance matching and corporate power combining to obtain the required output power levels. Losses in the power combining networks and impedance matching networks place limits on the maximum number of power combining stages N and the maximum impedance transformation ratio, hence the minimum feasible load resistance R.sub.L. The maximum power obtainable is then EQU P.sub.max =NV.sub.br.sup.2 /R.sub.L
The limit is severe. Transistor power amplifiers at microwave frequencies have relatively lower power, low bandwidth, and only moderate efficiency. Some 45 years after the invention of the transistor, vacuum electron devices (tubes) are still used heavily for microwave amplifiers in military radar systems.
What is needed is some type of innovative circuit topology that distributes the output voltage across many transistors, thus allowing output voltages several times the breakdown voltage of the transistors employed. Thus, large output power at microwave frequencies could be delivered into a large load with high efficiency. Prior attempts at this method have failed due to the method applied for sharing the total voltage between the several devices.